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 19-2682; Rev 0; 10/02
Notebook CPU Step-Down Controller
General Description
The MAX1718T step-down controller is intended for core CPU DC-to-DC converters in notebook computers. It features a dynamically adjustable output, ultra-fast transient response, high-DC accuracy, and high efficiency needed for leading-edge CPU core power supplies. Maxim's proprietary Quick-PWMTM quick-response, constant-ontime PWM control scheme handles wide input/output voltage ratios with ease and provides 100ns "instant-on" response to load transients while maintaining a relatively constant switching frequency. The output voltage can be dynamically adjusted through the 5-bit digital-to-analog converter (DAC) over a 0.6V to 1.75V range. The MAX1718T has an internal multiplexer that accepts three unique 5-bit VID DAC codes corresponding to performance, battery, and suspend modes. Precision slew-rate control provides "just-in-time" arrival at the new DAC setting, minimizing surge currents to and from the battery. A pair of complementary offset control inputs allows easy compensation for IR drops in PC board traces or creation of a voltage-positioned power supply. Voltage positioning modifies the load-transient response to reduce output capacitor requirements and total system power dissipation. Single-stage buck conversion allows these devices to directly step down high-voltage batteries for the highest possible efficiency. Alternatively, two-stage conversion (stepping down the 5V system supply instead of the battery) at a higher switching frequency allows the minimum possible physical size. The MAX1718T is available in a 28-pin QSOP package.
SHUTDOWN
Features
o o o o o o o o o o o o o o o o o Quick-PWM Architecture 1% VOUT Accuracy Over Line and Load 5-Bit On-Board DAC with Input Muxes Precision-Adjustable VOUT Slew Control 0.6V to 1.75V Output Adjust Range Precision Offset Control Supports Voltage-Positioned Applications 2V to 28V Battery Input Range Requires a Separate 5V Bias Supply 200kHz/300kHz/550kHz/1000kHz Switching Frequency Overvoltage/Undervoltage Protection Drives Large Synchronous-Rectifier FETs 700A (typ) ICC Supply Current 2A (typ) Shutdown Supply Current 2V 1% Reference Output VGATE Blanking During Transition Small 28-Pin QSOP Package
MAX1718T
Ordering Information
PART MAX1718TEEI TEMP RANGE -40C to +85C PIN-PACKAGE 28 QSOP
Minimal Operating Circuit
5V INPUT BATT 2V TO 28V
VCC SKP/SDN ILIM
VDD V+ BST
Applications
2-Cell to 4-Cell Li+ Battery to CPU Core Supply Converters 5V to CPU Core Supply Converters
D0 D1 DUAL MODETM VID MUX INPUTS D2 D3 D4
MAX1718T DH
LX DL GND
OUTPUT 0.6V TO 1.75V
MUX CONTROL SUSPEND INPUT DECODER
ZMODE SUS S0 S1 TIME FB NEG POS VCC
Pin Configuration appears at end of data sheet.
Patent pending.
CC REF TON
Quick-PWM is a trademark of Maxim Integrated Products, Inc.
VGATE OVP
POWER-GOOD OUTPUT
Dual Mode is a trademark of Maxim Integrated Products, Inc. ________________________________________________________________ Maxim Integrated Products 1
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
Notebook CPU Step-Down Controller MAX1718T
ABSOLUTE MAXIMUM RATINGS
V+ to GND ..............................................................-0.3V to +30V VCC, VDD to GND .....................................................-0.3V to +6V D0-D4, ZMODE, VGATE, OVP, SUS to GND ..........-0.3V to +6V SKP/SDN to GND ...................................................-0.3V to +16V ILIM, CC, REF, POS, NEG, S1, S0, TON, TIME to GND .....................................-0.3V to (VCC + 0.3V) DL to GND ..................................................-0.3V to (VDD + 0.3V) BST to GND ............................................................-0.3V to +36V DH to LX .....................................................-0.3V to (BST + 0.3V) LX to BST..................................................................-6V to +0.3V REF Short Circuit to GND ...........................................Continuous Continuous Power Dissipation 28-Pin QSOP (derate 10.8mW/C above +70C).........860mW Operating Temperature Range ..........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, V+ = 15V, VCC = VDD = SKP/SDN = 5V, VOUT = 1.25V, TA = 0C to +85C, unless otherwise noted.)
PARAMETER PWM CONTROLLER Input Voltage Range Battery voltage, V+ VCC, VDD V+ = 4.5V to 28V, includes load regulation error DAC codes from 0.9V to 1.75V DAC codes from 0.6V to 0.875V 2 4.5 -1 -1.5 5 -0.2 0.4 POS - NEG VFB / (POS - NEG); POS - NEG = 50mV 150kHz nominal, RTIME = 120k TIME Frequency Accuracy 380kHz nominal, RTIME = 47k 38kHz nominal, RTIME = 470k V+ = 5V, FB = 1.2V, TON = GND (1000kHz) On-Time (Note 1) TON = REF (550kHz) V+ = 12V, FB = 1.2V TON = open (300kHz) TON = VCC (200kHz) Minimum Off-Time (Note 1) BIAS AND REFERENCE Quiescent Supply Current (VCC) Quiescent Supply Current (VDD) Quiescent Battery Supply Current (V+) Shutdown Supply Current (VCC) Shutdown Supply Current (VDD) Shutdown Battery Supply Current (V+) SKP/SDN = GND SKP/SDN = GND SKP/SDN = GND, VCC = VDD = 0V or 5V Measured at VCC, FB forced above the regulation point Measured at VDD, FB forced above the regulation point 700 <1 25 2 <1 <1 1200 5 40 5 5 5 A A A A A A TON = VCC, open, or REF (200kHz, 300kHz, or 550kHz) TON = GND (1000kHz) -80 0.81 -8 -12 -12 230 165 320 465 260 190 355 515 400 300 0.86 +0.2 2.5 +80 0.91 +8 +12 +12 290 215 390 565 500 375 ns ns % 28 5.5 +1 % +1.5 mV A V mV V/V V CONDITIONS MIN TYP MAX UNITS
DC Output Voltage Accuracy Line Regulation Error Input Bias Current POS, NEG Common-Mode Range POS, NEG Differential Range POS, NEG Offset Gain
VCC = 4.5V to 5.5V, VBATT = 4.5V to 28V FB, POS, NEG
2
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Notebook CPU Step-Down Controller
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, V+ = 15V, VCC = VDD = SKP/SDN = 5V, VOUT = 1.25V, TA = 0C to +85C, unless otherwise noted.)
PARAMETER Reference Voltage Reference Load Regulation REF Sink Current FAULT PROTECTION Overvoltage Trip Threshold Overvoltage Fault Propagation Delay Output Undervoltage Fault Protection Threshold Output Undervoltage Fault Propagation Delay Output Undervoltage Fault Blanking Time Current-Limit Threshold Voltage (Positive, Default) Current-Limit Threshold Voltage (Positive, Adjustable) Current-Limit Threshold Voltage (Negative) Current-Limit Threshold Voltage (Zero Crossing) Current-Limit Default Switchover Threshold Thermal Shutdown Threshold VCC Undervoltage Lockout Threshold VGATE Lower Trip Threshold VGATE Upper Trip Threshold VGATE Propagation Delay VGATE Output Low Voltage VGATE Leakage Current GATE DRIVERS DH Gate-Driver On-Resistance DL Gate-Driver On-Resistance DH Gate-Driver Source/Sink Current DL Gate-Driver Sink Current DL Gate-Driver Source Current BST - LX forced to 5V DL, high state (pullup) DL, low state (pulldown) DH forced to 2.5V, BST - LX forced to 5V DL forced to 2.5V DL forced to 2.5V 1.0 1.0 0.4 1.6 4 1.6 3.5 3.5 1.0 A A A Hysteresis = +10C Rising edge, hysteresis = 20mV, PWM disabled below this level Measured at FB with respect to unloaded output voltage Measured at FB with respect to unloaded output voltage FB forced 2% outside VGATE trip threshold ISINK = 1mA High state, forced to 5.5V 4.1 -12 +8 -10 +10 10 0.4 1 Measured at FB FB forced 2% above trip threshold With respect to unloaded output voltage FB forced 2% below trip threshold From SKP/SDN signal going high, clock speed set by RTIME GND - LX, ILIM = VCC GND - LX LX - GND, ILIM = VCC GND - LX 3 TA = +25C to +85C TA = 0C to +85C ILIM = 0.5V ILIM = REF (2V) 90 85 35 165 -140 50 200 -117 4 VCC - 1 VCC - 0.4 150 4.4 -8 +12 65 1.95 2.00 10 70 10 256 100 110 115 65 230 -95 75 2.05 V s % s clks mV mV mV mV V C V % % s V A IREF = 0 to 50A REF in regulation 10 CONDITIONS VCC = 4.5V to 5.5V, no REF load MIN 1.98 TYP 2 MAX 2.02 0.01 UNITS V V A
MAX1718T
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3
Notebook CPU Step-Down Controller MAX1718T
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, V+ = 15V, VCC = VDD = SKP/SDN = 5V, VOUT = 1.25V, TA = 0C to +85C, unless otherwise noted.)
PARAMETER Dead Time LOGIC AND I/O Logic Input High Voltage Logic Input Low Voltage DAC B-Mode Programming Resistor, Low DAC B-Mode Programming Resistor, High D0-D4 Pullup/Pulldown Logic Input Current D0-D4, ZMODE, SUS, OVP D0-D4, ZMODE, SUS, OVP D0-D4, 0 to 0.4V or 2.6V to 5.5V applied through resistor, ZMODE = VCC D0-D4, 0 to 0.4V or 2.6V to 5.5V applied through resistor, ZMODE = VCC Entering impedance mode D0-D4, ZMODE = GND ZMODE, SUS, OVP For high Four Level Input Logic Levels (TON, S0, S1) SKP/SDN, S0, S1, and TON Input Current For open For REF For low SKP/SDN, S0, S1, TON forced to GND or VCC SKP/SDN = logic high (SKIP mode) SKP/SDN Input Levels SKP/SDN = open (PWM mode) SKP/SDN = logic low (shutdown mode) To enable no-fault mode SKP/SDN Float Level ISKP/SDN = 0A 12 1.8 -3 2.8 1.4 Pullup Pulldown -1 -1 VCC - 0.4 3.15 1.65 3.85 2.35 0.5 +3 6.0 2.2 0.5 15 2.2 V V A V 95 40 8 +1 +1 2.4 0.8 1.05 V V k k k A DL rising DH rising CONDITIONS MIN TYP 35 26 MAX UNITS ns
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, V+ = 15V, VCC = VDD = SKP/SDN = 5V, VOUT = 1.25V, TA = -40C to +85C, unless otherwise noted.) (Note 2)
PARAMETER PWM CONTROLLER DC Output Voltage Accuracy V+ = 4.5V to 28V, includes load regulation error DAC codes from 0.9V to 1.75V DAC codes from 0.6V to 0.875V -1.5 -2.0 -8 -12 -12 230 165 320 465 +1.5 % +2.0 +8 +12 +12 290 215 390 565 ns % CONDITIONS MIN TYP MAX UNITS
150kHz nominal, RTIME = 120k TIME Frequency Accuracy 380kHz nominal, RTIME = 47k 38kHz nominal, RTIME = 470k V+ = 5V, FB = 1.2V, TON = GND (1000kHz) On-Time (Note 1) TON = REF (550kHz) V+ = 12V, FB = 1.2V TON = open (300kHz) TON = VCC (200kHz)
4
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Notebook CPU Step-Down Controller
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, V+ = 15V, VCC = VDD = SKP/SDN = 5V, VOUT = 1.25V, TA = -40C to +85C, unless otherwise noted.) (Note 2)
PARAMETER Minimum Off-Time (Note 1) BIAS AND REFERENCE Quiescent Supply Current (VCC) Quiescent Supply Current (VDD) Quiescent Battery Supply Current (V+) Shutdown Supply Current (VCC) Shutdown Supply Current (VDD) Shutdown Battery Supply Current (V+) Reference Voltage FAULT PROTECTION Overvoltage Trip Threshold Output Undervoltage Protection Threshold Current-Limit Threshold Voltage (Positive, Default) Current-Limit Threshold Voltage (Positive, Adjustable) Current-Limit Threshold Voltage (Negative) VCC Undervoltage Lockout Threshold VGATE Lower Trip Threshold VGATE Upper Trip Threshold GATE DRIVERS DH Gate-Driver On-Resistance DL Gate-Driver On-Resistance LOGIC AND I/O Logic Input High Voltage Logic Input Low Voltage D0-D4, ZMODE, SUS, OVP D0-D4, ZMODE, SUS, OVP 2.4 0.8 V V BST - LX forced to 5V DL, high state (pullup) DL, low state (pulldown) 3.5 3.5 1.0 Measured at FB With respect to unloaded output voltage GND - LX, ILIM = VCC GND - LX LX - GND, ILIM = VCC Rising edge, hysteresis = 20mV, PWM disabled below this level Measured at FB with respect to unloaded output voltage Measured at FB with respect to unloaded output voltage ILIM = 0.5V ILIM = REF (2V) 1.95 65 80 33 160 -145 4.1 -12.5 +7.5 2.05 75 115 65 240 -90 4.4 -7.5 +12.5 V % mV mV mV V % % SKP/SDN = 0 SKP/SDN = 0 SKP/SDN = 0, VCC = VDD = 0 or 5V VCC = 4.5V to 5.5V, no REF load 1.98 Measured at VCC, FB forced above the regulation point Measured at VDD, FB forced above the regulation point 1300 5 40 5 5 5 2.02 A A A A A A V CONDITIONS TON = VCC, open, or REF (200kHz, 300kHz, or 550kHz) TON = GND (1000kHz) MIN TYP MAX 500 375 UNITS ns
MAX1718T
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5
Notebook CPU Step-Down Controller MAX1718T
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, V+ = 15V, VCC = VDD = SKP/SDN = 5V, VOUT = 1.25V, TA = -40C to +85C, unless otherwise noted.) (Note 2)
PARAMETER DAC B-Mode Programming Resistor, Low DAC B-Mode Programming Resistor, High CONDITIONS D0-D4, 0 to 0.4V or 2.6V to 5.5V applied through resistor, ZMODE = VCC D0-D4, 0 to 0.4V or 2.6V to 5.5V applied through resistor, ZMODE = VCC 95 MIN TYP MAX 1.05 UNITS k k
Note 1: On-time specifications are measured from 50% to 50% at the DH pin, with LX forced to 0V, BST forced to 5V, and a 500pF capacitor from DH to LX to simulate external MOSFET gate capacitance. Actual in-circuit times can be different due to MOSFET switching speeds. Note 2: Specifications to TA = -40C are guaranteed by design and not production tested.
Typical Operating Characteristics
(Circuit of Figure 1, V+ = 12V, VDD = VCC = SKP/SDN = 5V, VOUT = 1.25V, TA = +25C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT 300kHz VOLTAGE POSITIONED
MAX1718T toc01
FREQUENCY vs. LOAD CURRENT
MAX1718T toc02
FREQUENCY vs. INPUT VOLTAGE
340 330 FREQUENCY (kHz) 320 310 300 290 280 IOUT = 3A IOUT = 18A
MAX1718T toc03
95 90 85 EFFICIENCY (%) 80 75 70 65 60 55 50 0.01 0.1 1 LOAD CURRENT (A) 10 SKIP MODE V+ = 12V SKIP MODE V+ = 20V PWM MODE V+ = 7V PWM MODE V+ = 12V PWM MODE V+ = 20V SKIP MODE V+ = 7V
400 PWM MODE 300 FREQUENCY (kHz) SKIP MODE 200
350
100
270 260
0 100 0 5 10 LOAD CURRENT (A) 15 20
250 7.0 10.4 13.8 17.2 20.6 24.0 INPUT VOLTAGE (V)
FREQUENCY vs. TEMPERATURE
MAX1718T toc04
OUTPUT CURRENT AT CURRENT LIMIT vs. TEMPERATURE
MAX1718T toc05
NO-LOAD SUPPLY CURRENT vs. INPUT VOLTAGE
900 800 SUPPLY CURRENT (A) 700 600 500 400 300 200 100 I+ 5 10 15 INPUT VOLTAGE (V) 20 25 ICC + IDD
MAX1718T toc06
326.0 325.5 FREQUENCY (kHz) 325.0 324.5 324.0 323.5 323.0
IOUT = 19A
50 45 40 35 30 25 20
1000
CURRENT (A)
-40
-15
10
35
60
85
0 -40 -15 10 35 60 85 TEMPERATURE (C)
TEMPERATURE (C)
6
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Notebook CPU Step-Down Controller
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V+ = 12V, VDD = VCC = SKP/SDN = 5V, VOUT = 1.25V, TA = +25C, unless otherwise noted.)
NO-LOAD SUPPLY CURRENT vs. INPUT VOLTAGE
35 SUPPLY CURRENT (mA) 30 25 20 ICC + IDD 15 I+ 10 5 0 5 10 15 INPUT VOLTAGE (V) 20 25 40s/div A = VOUT, 50mV/div, AC-COUPLED B = INDUCTOR CURRENT, 10A/div 40s/div A = VOUT, 50mV/div, AC-COUPLED B = INDUCTOR CURRENT, 10A/div B 0A B 0A
MAX1718T toc07
MAX1718T
LOAD-TRANSIENT RESPONSE (SKIP MODE)
MAX1718T toc08a
LOAD-TRANSIENT RESPONSE (PWM MODE)
MAX1718T toc08b
40
A
A
STARTUP WAVEFORM (PWM MODE, NO LOAD)
MAX1718T toc09
STARTUP WAVEFORM (PWM MODE, IOUT = 12A)
DYNAMIC OUTPUT VOLTAGE TRANSITION (PWM MODE)
MAX1718T toc11
MAX1718T toc10
A
A
A
B B 0A B 0A
0A
C
C
C D 100s/div A = VOUT, 1V/div B = INDUCTOR CURRENT, 10A/div C = SKP/SDN, 5V/div 40s/div VOUT = 1.15V TO 1.25V IOUT = 3A, RTIME = 62k A = VOUT, 100mV/div, AC-COUPLED B = INDUCTOR CURRENT, 10A/div C = VGATE, 5V/div D = ZMODE, 5V/div
100s/div A = VOUT, 1V/div B = INDUCTOR CURRENT, 10A/div C = SKP/SDN, 5V/div
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7
Notebook CPU Step-Down Controller MAX1718T
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V+ = 12V, VDD = VCC = SKP/SDN = 5V, VOUT = 1.25V, TA = +25C, unless otherwise noted.)
DYNAMIC OUTPUT VOLTAGE TRANSITION (PWM MODE)
MAX1718T toc12
SHUTDOWN WAVEFORM (PWM MODE, NO LOAD)
MAX1718T toc13
SHUTDOWN WAVEFORM (PWM MODE, IOUT = 12A)
MAX1718T toc14
A A A
B
0A
B
0A B 0A
C D 40s/div VOUT = 0.7V TO 1.25V IOUT = 3A, RTIME = 62k A = VOUT, 500mV/div, AC-COUPLED B = INDUCTOR CURRENT, 10A/div C = VGATE, 5V/div D = SUS, 5V/div
C
C 100s/div A = VOUT, 1V/div B = INDUCTOR CURRENT, 10A/div C = SKP/SDN, 5V/div 100s/div A = VOUT, 1V/div B = INDUCTOR CURRENT, 10A/div C = SKP/SDN, 5V/div
OFFSET FUNCTION SCALE FACTOR vs. DAC SETTING
MAX1718T toc15
OUTPUT VOLTAGE vs. POS-NEG DIFFERENTIAL
1.40 1.35 OUTPUT VOLTAGE (V) 1.30 1.25 1.20 1.15 1.10
MAX1718T toc16
0.925 0.900 POS-NEG SCALE FACTOR 0.875 0.850 0.825 0.800 0.775 0.750 0.5 0.7 0.9 1.1 1.3 1.5 1.7 THEORETICAL MEASURED
1.45
1.05 1.00 1.9 -300 -200 -100 0 100 200 DAC SETTING (V) POS-NEG (V)
8
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Notebook CPU Step-Down Controller MAX1718T
Typical Operating Characteristics (continued)
(Circuit of Figure 1, V+ = 12V, VDD = VCC = SKP/SDN = 5V, VOUT = 1.25V, TA = +25C, unless otherwise noted.)
OUTPUT VOLTAGE DISTRIBUTION
MAX1718T toc17
REFERENCE VOLTAGE DISTRIBUTION
MAX1718T toc18
25 VOUT = 1.25V SAMPLE PERCENTAGE (%) 20
25
15
SAMPLE PERCENTAGE (%)
20
15
10
10
5
5
0 -0.48
-0.24
0
0.24
0.48
0 1.995
1.998
2.000
2.002
2.005
OUTPUT VOLTAGE ERROR (%)
REFERENCE VOLTAGE (V)
Pin Description
PIN 1 NAME V+ FUNCTION Battery Voltage Sense Connection. Connect V+ to input power source. V+ is used only for PWM one-shot timing. DH on-time is inversely proportional to input voltage over a range of 2V to 28V.
2
Combined Shutdown and Skip-Mode Control. Drive SKP/SDN to GND for shutdown. Leave SKP/SDN open for low-noise forced-PWM mode, or drive to VCC for pulse-skipping operation. Low-noise forced-PWM mode SKP/SDN causes inductor current recirculation at light loads and suppresses pulse-skipping operation. Forcing SKP/SDN to 12V to 15V disables both the overvoltage protection and undervoltage protection circuits and clears the fault latch, with otherwise normal pulse-skipping operation. Do not connect SKP/SDN to > 15V. TIME FB NEG Slew-Rate Adjustment Pin. Connect a resistor from TIME to GND to set the internal slew-rate clock. A 470k to 47k resistor sets the clock from 38kHz to 380kHz, fSLEW = 150kHz 5 120k / RTIME. Feedback Input. Connect FB to the junction of the external inductor and the positioning resistor (Figure 1). Feedback Offset Adjust Negative Input. The output shifts by an amount equal to the difference between POS and NEG multiplied by a scale factor that depends on the DAC codes (see the Integrator Amplifiers/Output Voltage Offsets section). Connect both POS and NEG to REF if the offset function is not used. Integrator Capacitor Connection. Connect a 47pF to 1000pF (47pF typ) capacitor from CC to GND to set the integration time constant (see the Integrator Amplifiers/Output Voltage Offsets section). Suspend-Mode Voltage Select Input. S0 and S1 are four-level digital inputs that select the suspend-mode VID code for the suspend-mode multiplexer inputs. If SUS is high, the suspend-mode VID code is delivered to the DAC (see the Internal Multiplexers (ZMODE/SUS) section). Analog Supply Voltage Input for PWM Core. Connect VCC to the system supply voltage (4.5V to 5.5V) with a series 20 resistor. Bypass to GND with a 0.22F (min) capacitor.
3 4 5
6
CC
7, 8
S0, S1
9
VCC
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9
Notebook CPU Step-Down Controller MAX1718T
Pin Description (continued)
PIN NAME FUNCTION On-Time Selection Control Input. This is a four-level input that sets the K factor (Table 2) to determine DH on-time. Connect TON to the following pins for the indicated operation: GND = 1000kHz REF = 550kHz Open = 300kHz VCC = 200kHz 2V Reference Output. Bypass to GND with 0.22F (min) capacitor. Can source 50A for external loads. Loading REF degrades FB accuracy according to the REF load-regulation error. Current-Limit Adjustment. The GND - LX current-limit threshold defaults to 100mV if ILIM is connected to VCC. In adjustable mode, the current-limit threshold voltage is 1/10th the voltage seen at ILIM over a 0.5V to 3V range. The logic threshold for switchover to the 100mV default value is approximately VCC - 1V. Connect ILIM to REF for a fixed 200mV threshold. Feedback Offset Adjust Negative Input. The output shifts by an amount equal to the difference between POS and NEG multiplied by a scale factor that depends on the DAC codes (see the Integrator Amplifiers/Output Voltage Offsets section). Connect both POS and NEG to REF if the offset function is not used. Open-Drain Power-Good Output. VGATE is normally high when the output is in regulation. If VFB is not within a 10% window of the DAC setting, VGATE is asserted low. During DAC code transitions, VGATE is forced high until 1 clock period after the slew-rate controller finishes the transition. VGATE is low during shutdown. Analog and Power Ground. Also connects to the current-limit comparator. Low-Side Gate-Driver Output. DL swings GND to VDD. Supply Voltage Input for the DL Gate Driver, 4.5V to 5.5V. Bypass to GND with a 1F capacitor. Suspend-Mode Control Input. When SUS is high, the suspend-mode VID code, as programmed by S0 and S1, is delivered to the DAC. Connect SUS to GND if the suspend-mode multiplexer is not used (see the Internal Multiplexers (ZMODE/SUS) section). Performance-Mode MUX Control Input. If SUS is low, ZMODE selects between two different VID DAC codes. If ZMODE is low, the VID DAC code is set by the logic-level voltages on D0-D4. On the rising edge of ZMODE, during power-up with ZMODE high, or on the falling edge of SUS when ZMODE is high, the VID DAC code is determined by the impedance at D0-D4 (see the Internal Multiplexers (ZMODE/SUS) section). Overvoltage Protection Control Input. Connect OVP low to enable overvoltage protection. Connect OVP high to disable overvoltage protection. The overvoltage trip threshold is approximately 2V. The state of OVP does not affect output undervoltage fault protection or thermal shutdown. VID DAC Code Inputs. D0 is the LSB, and D4 is the MSB of the internal 5-bit VID DAC (Table 3). If ZMODE is low, D0-D4 are high-impedance digital inputs, and the VID DAC code is set by the logic-level voltages on D0-D4. On the rising edge of ZMODE, during power-up with ZMODE high, or on the falling edge of SUS when ZMODE is high, the VID DAC code is determined by the impedance at D0-D4 as follows: Logic low = source impedance is 1k + 5%. Logic high = source impedance is 100k - 5%. Boost Flying Capacitor Connection. Connect BST to the external boost diode and capacitor as shown in Figure 1. An optional resistor in series with BST allows the DH pullup current to be adjusted (Figure 8). Inductor Connection. LX is the internal lower supply rail for the DH high-side gate driver. It also connects to the current-limit comparator and the skip-mode zero-crossing comparator. High-Side Gate-Driver Output. DH swings LX to BST.
10
TON
11
REF
12
ILIM
13
POS
14 15 16 17 18
VGATE GND DL VDD SUS
19
ZMODE
20
OVP
21-25
D4-D0
26 27 28
BST LX DH
10
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Notebook CPU Step-Down Controller MAX1718T
R1 20 C7 1F 2 10 25 24 R2 100k R3 100k 23 9 VCC SKP/SDN TON D0 D1 D2 LX 27 17 VDD V+ BST 1 26
5V INPUT C1 1F
BATT 7V TO 24V C2, 25V, X5R 5 x 10F
SHUTDOWN VCC
D1 CMPSH-3 2x IRF7811A Q1 C3 0.1F 2x L1 0.68H SUMIDA CEP125#4712-TO11 D2 CENTRAL SEMICONDUCTOR CMSH5-40
DH
28
R8 0.004
OUTPUT 0.6V TO 1.75V C4 6 x 270F, 2V PANASONIC SP EEFUE0D271R
22 21
MAX1718T
D3 D4 GND DL
16 15 4
FDS7764A Q2
MUX CONTROL REF SUSPEND INPUT DECODER R4 62k C6 47pF
19 ZMODE 18 SUS 7 S0 8 S1 3
FB
NEG
5
TIME 13 R6 511k REF VGATE 14 20 R7 4.75k 5V R5 100k POWER-GOOD OUTPUT
6 C5 0.22F 11 R18 24.9k R19 27.4k
CC
POS
12
ILIM
OVP
Figure 1. Standard Application Circuit
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11
Notebook CPU Step-Down Controller MAX1718T
Table 1. Component Suppliers
MANUFACTURER Central Semiconductor Dale-Vishay Fairchild International Rectifier Kemet Motorola Nihon Panasonic Sanyo SGS-Thomson Sumida Taiyo Yuden TDK Toko Zetex PHONE 516-435-1110 402-564-3131 408-721-2181 310-322-3331 408-986-0424 602-303-5454 847-843-7500 714-373-7939 619-661-6835 617-259-0300 708-956-0666 408-573-4150 847-390-4373 800-745-8656 516-543-7100 FAX 516-435-1824 402-563-6418 408-721-1635 310-322-3332 408-986-1442 602-994-6430 847-843-2798 714-373-7183 619-661-1055 617-259-9442 708-956-0702 408-573-4159 847-390-4428 408-943-9790 516-864-7630
Free-Running, Constant-On-Time PWM Controller with Input Feed Forward
The Quick-PWM control architecture is a pseudofixedfrequency, constant-on-time current-mode type with voltage feed forward (Figure 2). This architecture relies on the output filter capacitor's ESR to act as the current-sense resistor, so the output ripple voltage provides the PWM ramp signal. The control algorithm is simple: the high-side switch on-time is determined solely by a one-shot whose period is inversely proportional to input voltage and directly proportional to output voltage. Another one-shot sets a minimum off-time (400ns typ). The on-time one-shot is triggered if the error comparator is low, the low-side switch current is below the current-limit threshold, and the minimum off-time oneshot has timed out.
On-Time One-Shot (TON)
The heart of the PWM core is the one-shot that sets the high-side switch on-time. This fast, low-jitter, adjustable one-shot includes circuitry that varies the on-time in response to battery and output voltage. The high-side switch on-time is inversely proportional to the battery voltage as measured by the V+ input, and proportional to the output voltage. This algorithm results in a nearly constant switching frequency despite the lack of a fixed-frequency clock generator. The benefits of a constant switching frequency are twofold: first, the frequency can be selected to avoid noise-sensitive regions such as the 455kHz IF band; second, the inductor ripple-current operating point remains relatively constant, resulting in easy design methodology and predictable output voltage ripple: On-time = K (VOUT + 0.075V) / VIN where K is set by the TON pin-strap connection and 0.075V is an approximation to accommodate the expected drop across the low-side MOSFET switch (Table 2). The on-time one-shot has good accuracy at the operating points specified in the Electrical Characteristics table (10% at 200kHz and 300kHz, 12% at 550kHz and 1000kHz). On-times at operating points far removed from the conditions specified in the Electrical Characteristics table can vary over a wider range. For example, the 1000kHz setting typically runs about 10% slower with inputs much greater than +5V due to the very short ontimes required. On-times translate only roughly to switching frequencies. The on-times guaranteed in the Electrical Characteristics table are influenced by switching delays in the
Detailed Description
5V Bias Supply (VCC and VDD) The MAX1718T requires an external 5V bias supply in addition to the battery. Typically, this 5V bias supply is the notebook's 95%-efficient 5V system supply. Keeping the bias supply external to the IC improves efficiency and eliminates the cost associated with the 5V linear regulator that would otherwise be needed to supply the PWM circuit and gate drivers. If stand-alone capability is needed, the 5V supply can be generated with an external linear regulator. The 5V bias supply must provide VCC (PWM controller) and VDD (gate-drive power), so the maximum current drawn is:
IBIAS = ICC + f (QG1 + QG2) = 10mA to 40mA (typ) where ICC is 800A (typ), f is the switching frequency, and QG1 and QG2 are the MOSFET data sheet total gate-charge specification limits at VGS = 5V. V+ and VDD can be tied together if the input power source is a fixed 4.5V to 5.5V supply. If the 5V bias supply is powered up prior to the battery supply, the enable signal (SKP/SDN going from low to high or open) must be delayed until the battery voltage is present to ensure startup.
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Notebook CPU Step-Down Controller MAX1718T
VBATT 2V TO 28V REF
V+
ILIM
MAX1718T
TOFF TON ON-TIME COMPUTE FROM D/A 1-SHOT TRIG Q 5V 9
1
BST
TON TRIG 1-SHOT Q
S R
Q CURRENT LIMIT DH LX
SKP/SDN
ERROR AMP REF 70k 10k ZERO CROSSING VDD DL S Q gm gm R GND FB REF -10% REF +10% OUTPUT 5V
CC
REF
NEG POS FB OVP/UVP DETECT CHIP SUPPLY
VCC
5V
VGATE
R-2R D/A CONVERTER
2V REF
REF
OVP ZMODE
MUXES AND SLEW CONTROL SUS S1 S0 D0 D1 D2 D3 D4 TIME
Figure 2. Functional Diagram
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Notebook CPU Step-Down Controller MAX1718T
Table 2. Approximate K-Factor Errors
TON SETTING VCC OPEN REF GND TON FREQUENCY (kHZ) 200 300 550 1000 K-FACTOR (s) 5 3.3 1.8 1.0 APPROXIMATE K-FACTOR ERROR (%) 10 10 12.5 12.5 MIN RECOMMENDED VBATT AT VOUT = 1.25V (V) 1.7 1.8 2.6 3.6 VOUT = 1.75V (V) 2.3 2.5 3.5 4.9
external high-side MOSFET. Resistive losses, including the inductor, both MOSFETs, output capacitor ESR, and PC board copper losses in the output and ground tend to raise the switching frequency at higher output currents. Also, the dead-time effect increases the effective on-time, reducing the switching frequency. It occurs only in PWM mode (SKP/SDN = open) and during dynamic output voltage transitions when the inductor current reverses at light or negative load currents. With reversed inductor current, the inductor's EMF causes LX to go high earlier than normal, extending the on-time by a period equal to the DH-rising dead time. For loads above the critical conduction point, where the dead-time effect is no longer a factor, the actual switching frequency is: f= (VOUT + VDROP1) tON (VIN + VDROP1 - VDROP2 )
where VDROP1 is the sum of the parasitic voltage drops in the inductor discharge path, including synchronous rectifier, inductor, and PC board resistances; VDROP2 is the sum of the parasitic voltage drops in the inductor charge path, including high-side switch, inductor, and PC board resistances; and tON is the on-time calculated by the MAX1718T.
at the CC pin. Use a capacitor value of 47pF to 1000pF (47pF typ). The gm of each amplifier is 160S (typ). The POS/NEG amplifier is used to add small offsets to the VID DAC setting or to correct for voltage drops. To create an output offset, bias POS and NEG to a voltage (typically V OUT or REF) within their common-mode range, and offset them from one another with a resistive divider (Figures 3 and 4). If VPOS is higher than VNEG, then the output is shifted in the positive direction. If VNEG is higher than VPOS, then the output is shifted in the negative direction. The amount of output offset is less than the difference from POS to NEG by a scale factor that varies with the VID DAC setting as shown in Table 3. The common-mode range of POS and NEG is 0.4V to 2.5V. For applications that require multiple offsets, an external multiplexer can be used to select various resistor values (Figure 5). Both the integrator amplifiers can be disabled by connecting NEG to VCC.
Forced-PWM Mode (SKP/SDN Open)
The low-noise forced-PWM mode (SKP/SDN open) disables the zero-crossing comparator, allowing the inductor current to reverse at light loads. This causes the low-side gate-drive waveform to become the complement of the high-side gate-drive waveform. The benefit of forced-PWM mode is to keep the switching frequency fairly constant, but it comes at a cost: The no-load battery current can be 10mA to 40mA, depending on the external MOSFETs and switching frequency. Forced-PWM mode is required during downward output voltage transitions. The MAX1718T uses PWM mode during all transitions, but only while the slew-rate controller is active. Due to voltage positioning, when a transition uses high negative inductor current, the output voltage does not settle to its final intended value until well after the slew-rate controller terminates. Consequently it is possible, at very high negative slew currents, for the output to end up high enough to cause VGATE to go low.
Integrator Amplifiers/Output Voltage Offsets
Two transconductance integrator amplifiers provide a fine adjustment to the output regulation point. One amplifier forces the DC average of the feedback voltage to equal the VID DAC setting. The second amplifier is used to create small positive or negative offsets from the VID DAC setting, using the POS and NEG pins. The integrator block has the ability to lower the output voltage by 8% and raise it by 8%. For each amplifier, the differential input voltage range is at least 80mV total, including DC offset and AC ripple. The two amplifiers' outputs are directly summed inside the chip, so the integration time constant can be set easily with one capacitor
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Notebook CPU Step-Down Controller
Thus, it is necessary to use forced-PWM mode during all negative transitions. Most applications should use PWM mode exclusively, although there is some benefit to using skip mode while in the low-power suspend state (see the Using Skip Mode During Suspend (SKP/SDN = VCC) section.)
MAX1718T
REF
MAX1718T POS
Automatic Pulse-Skipping Switchover
NEG
Figure 3. Resistive Divider from REF
DH
In skip mode (SKP/SDN high), an inherent automatic switchover to PFM takes place at light loads (Figure 6). This switchover is effected by a comparator that truncates the low-side switch on-time at the inductor current's zero crossing. This mechanism causes the threshold between pulse-skipping PFM and nonskipping PWM operation to coincide with the boundary between continuous and discontinuous inductor-current operation. The load-current level at which PFM/PWM crossover occurs, ILOAD(SKIP), is equal to 1/2 the peak-to-peak ripple current, which is a function of the inductor value (Figure 6). For a battery range of 7V to 24V, this threshold is relatively constant, with only a minor dependence on battery voltage: I LOAD(SKIP) K x VOUT 2xL x VBATT - VOUT VBATT
DL MAX1718T POS
NEG
where K is the on-time scale factor (Table 2). For example, in the standard application circuit this becomes: 3.3s x 1.25V 12V - 1.25V x = 2.7A 2 x 0.68H 12V
Figure 4. Resistive Divider from OUTPUT
The crossover point occurs at a lower value if a swinging (soft-saturation) inductor is used. The switching waveforms may appear noisy and asynchronous when light loading causes pulse-skipping operation, but this is a normal operating condition that results in high light-load efficiency. Trade-offs in PFM noise vs. light-load efficiency are made by varying the inductor value. Generally, low inductor values produce a broader efficiency vs. load curve, while higher values result in higher full-load efficiency (assuming that the coil resistance remains fixed) and less output voltage ripple. Penalties for using higher inductor values include larger physical size and degraded load-transient response, especially at low input voltage levels.
DH
DL MAX1718T POS
NEG MUX MAX4524 SEL A B
Current-Limit Circuit
The current-limit circuit employs a unique "valley" currentsensing algorithm that uses the on-resistance of the low-side MOSFET as a current-sensing element. If the current-sense signal is above the current-limit threshold, the PWM is not allowed to initiate a new cycle (Figure 7). The actual peak current is greater than the
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Figure 5. Programmable Offset Voltage
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Notebook CPU Step-Down Controller MAX1718T
i t INDUCTOR CURRENT
=
VBATT - VOUT L
IPEAK IPEAK ILOAD INDUCTOR CURRENT
ILOAD = IPEAK/2
ILIMIT
0
ON-TIME
TIME
0
TIME
Figure 6. Pulse-Skipping/Discontinuous Crossover Point
Figure 7. "Valley" Current-Limit Threshold Point
current-limit threshold by an amount equal to the inductor ripple current. Therefore, the exact current-limit characteristic and maximum load capability are a function of the MOSFET on-resistance, inductor value, and battery voltage. The reward for this uncertainty is robust, lossless overcurrent sensing. When combined with the undervoltage protection circuit, this currentlimit method is effective in almost every circumstance. There is also a negative current limit that prevents excessive reverse inductor currents when VOUT is sinking current. The negative current-limit threshold is set to approximately 120% of the positive current limit, and therefore tracks the positive current limit when ILIM is adjusted. The current-limit threshold is adjusted with an external resistor-divider at ILIM. The current-limit threshold voltage adjustment range is from 50mV to 300mV. In the adjustable mode, the current-limit threshold voltage is precisely 1/10th the voltage seen at ILIM. The threshold defaults to 100mV when ILIM is connected to VCC. The logic threshold for switchover to the 100mV default value is approximately VCC - 1V. The adjustable current limit accommodates MOSFETs with a wide range of on-resistance characteristics (see the Design Procedure section). For a high-accuracy current-limit application, see Figure 16. Carefully observe the PC board layout guidelines to ensure that noise and DC errors do not corrupt the current-sense signals seen by LX and GND. Place the IC close to the low-side MOSFET with short, direct traces, making a Kelvin-sense connection to the source and drain terminals.
MOSFET Gate Drivers (DH, DL)
The DH and DL drivers are optimized for driving moderate-sized high-side and larger low-side power MOSFETs. This is consistent with the low duty factor seen in the notebook CPU environment, where a large VBATT - VOUT differential exists. An adaptive dead-time circuit monitors the DL output and prevents the highside FET from turning on until DL is fully off. There must be a low-resistance, low-inductance path from the DL driver to the MOSFET gate for the adaptive dead-time circuit to work properly. Otherwise, the sense circuitry in the MAX1718T interprets the MOSFET gate as "off" while there is actually still charge left on the gate. Use very short, wide traces measuring 10 to 20 squares (50 mils to 100 mils wide if the MOSFET is 1in from the MAX1718T). The dead time at the other edge (DH turning off) is determined by a fixed 35ns (typ) internal delay. The internal pulldown transistor that drives DL low is robust, with a 0.4 (typ) on-resistance. This helps prevent DL from being pulled up during the fast rise time of the inductor node, due to capacitive coupling from the drain to the gate of the low-side synchronous-rectifier MOSFET. However, for high-current applications, you might still encounter some combinations of high- and low-side FETs that causes excessive gate-drain coupling, which can lead to efficiency-killing, EMIproducing shoot-through currents. This is often remedied by adding a resistor in series with BST, which increases the turn-on time of the high-side FET without degrading the turn-off time (Figure 8).
POR
Power-on reset (POR) occurs when VCC rises above approximately 2V, resetting the fault latch and preparing the PWM for operation. V CC undervoltage lockout (UVLO) circuitry inhibits switching, forces VGATE low,
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Notebook CPU Step-Down Controller
clamped with a Schottky diode to GND to reduce the negative excursion.
VBATT
MAX1718T
+5V
DAC Inputs D0-D4
BST 5 TYP
DH LX
MAX1718T
Figure 8. Reducing the Switching-Node Rise Time
and forces the DL gate driver high (to enforce output overvoltage protection). When VCC rises above 4.2V, the DAC inputs are sampled and the output voltage begins to slew to the DAC setting. For automatic startup, the battery voltage should be present before VCC. If the MAX1718T attempts to bring the output into regulation without the battery voltage present, the fault latch trips. The SKP/SDN pin can be toggled to reset the fault latch.
The DAC programs the output voltage. It typically receives a preset digital code from the CPU pins, which are either hardwired to GND or left open circuit. They can also be driven by digital logic, general-purpose I/O, or an external mux. Do not leave D0-D4 floating--use 1M or less pullups if the inputs may float. D0-D4 can be changed while the SMPS is active, initiating a transition to a new output voltage level. If this mode of DAC control is used, connect ZMODE and SUS low. Change D0-D4 together, avoiding greater than 1s skew between bits. Otherwise, incorrect DAC readings may cause a partial transition to the wrong voltage level, followed by the intended transition to the correct voltage level, lengthening the overall transition time. The available DAC codes and resulting output voltages are listed in Table 3.
Internal Multiplexers (ZMODE, SUS)
The MAX1718T has two unique internal VID input multiplexers (muxes) that can select one of three different VID DAC code settings for different processor states. Depending on the logic level at SUS, the suspend (SUS) mode mux selects the VID DAC code settings from either the ZMODE mux or the S0/S1 input decoder. The ZMODE mux selects one of the two VID DAC code settings from the D0-D4 pins, based on either voltage on the pins or the output of the impedance decoder (Figure 9). When SUS is high, the suspend mode mux selects the VID DAC code settings from the S0/S1 input decoder. The outputs of the decoder are determined by inputs S0 and S1 (Table 4). When SUS is low, the suspend mode mux selects the output of the ZMODE mux. Depending on the logic level at ZMODE, the ZMODE mux selects the VID DAC code settings using either the voltage on D0-D4 or the output of the impedance decoder (Table 5). If ZMODE is low, the logic-level voltages on D0-D4 set the VID DAC settings. This is called logic mode. In this mode, the inputs are continuously active and can be dynamically changed by external logic. The logic mode VID DAC code setting is typically used for the battery mode state, and the source of this code is sometimes the VID pins of the CPU with suitable pullup resistors. On the rising edge of ZMODE, during power-up with ZMODE high or on the falling edge of SUS when ZMODE is high, the impedances at D0-D4 are sampled by the impedance decoder to see if a large resistance
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Shutdown
When SKP/SDN goes low, the MAX1718T enters lowpower shutdown mode. VGATE goes low immediately. The output voltage ramps down to 0V in 25mV steps at the clock rate set by RTIME. When the DAC reaches the 0V setting, DL goes high, DH goes low, the reference is turned off, and the supply current drops to about 2A. When SKP/SDN goes high or floats, the reference powers up, and after the reference UVLO is passed, the DAC target is evaluated and switching begins. The slew-rate controller ramps up from 0V in 25mV steps to the currently selected code value (based on ZMODE and SUS). There is no traditional soft-start (variable current limit) circuitry, so full output current is available immediately. VGATE goes high after the slew-rate controller has terminated and the output voltage is in regulation.
UVLO
If VCC drops low enough to trip the UVLO comparator, it is assumed that there is not enough supply voltage to make valid decisions. To protect the output from overvoltage faults, DL is forced high in this mode. This forces the output to GND, but it does not use the slewrate controller. This results in large negative inductor current and possibly small negative output voltages. If VCC is likely to drop in this fashion, the output can be
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Notebook CPU Step-Down Controller MAX1718T
Table 3. Output Voltage vs. DAC Codes
D4 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 D3 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 D2 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 D1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 D0 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 OUTPUT VOLTAGE (V) 1.75 1.70 1.65 1.60 1.55 1.50 1.45 1.40 1.35 1.30 1.25 1.20 1.15 1.10 1.05 1.00 0.975 0.950 0.925 0.900 0.875 0.850 0.825 0.800 0.775 0.750 0.725 0.700 0.675 0.650 0.625 0.600 POS/NEG SCALE FACTOR 0.90 0.90 0.90 0.89 0.89 0.89 0.88 0.88 0.88 0.87 0.87 0.86 0.86 0.85 0.85 0.84 0.84 0.83 0.83 0.82 0.82 0.82 0.81 0.81 0.80 0.80 0.79 0.78 0.78 0.77 0.76 0.76
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Notebook CPU Step-Down Controller
is in series with the pin. This is called Impedance mode. If the voltage level on the pin is a logic low, an internal switch connects the pin to an internal 26k pullup for about 4s to see if the pin voltage can be forced high (Figure 10). If the pin voltage can be pulled to a logic high, the impedance is considered high and so is the Impedance mode logic state. Similarly, if the voltage level on the pin is a logic high, an internal switch connects the pin to an internal 8k pulldown to see if the pin voltage can be forced low. If so, the pin is high impedance and its impedance mode logic state is high. In either sampling condition, if the pin's logic level does not change, the pin is determined to be low impedance and the impedance mode logic state is low. A high pin impedance (and logic high) is 100k or greater, and a low impedance (and logic low) is 1k or less. The Electrical Characteristics table guaranteed levels for these impedances are 95k and 1.05k to allow the use of standard 100k and 1k resistors with 5% tolerance.
MAX1718T
Table 4. Suspend Mode DAC Codes
S1 GND GND GND GND REF REF REF REF OPEN OPEN OPEN OPEN VCC VCC VCC VCC S0 GND REF OPEN VCC GND REF OPEN VCC GND REF OPEN VCC GND REF OPEN VCC OUTPUT VOLTAGE (V) 0.975 0.950 0.925 0.900 0.875 0.850 0.825 0.800 0.775 0.750 0.725 0.700 0.675 0.650 0.625 0.600
Using the ZMODE Mux
There are many ways to use the versatile ZMODE mux. The preferred method depends on when and how the VID DAC codes for the various states are determined. If the output voltage codes are fixed at PC board design time, program both codes with a simple combination of pin-strap connections and series resistors (Figure 11). If the output voltage codes are chosen during PC board assembly, both codes can be independently programmed with resistors (Figure 12). This matrix of 10 resistor-footprints can be programmed to all possible logic mode and impedance mode code combinations with only five resistors.
Often the CPU pins provide one set of codes that are typically used with pullup resistors to provide the logic mode VID code, and resistors in series with D0-D4 set the impedance mode code. Since some of the CPU's VID pins may float, the open-circuit pins can present a problem for the ZMODE mux's impedance mode. For the impedance mode to work, any pins intended to be low during impedance mode must appear to be low impedance, at least for the 4s sampling interval. This can be achieved in several ways, including the following two (Figure 13). By using low-impedance pullup
D0 D1 D2 D3 D4 IMPEDANCE DECODER IN VCC POR ZMODE S0 S1 SUS
ZMODE MUX 1 OUT 0 SEL OUT S0/S1 DECODER OUT IN DAC 0 SUS MUX
1 SEL
Figure 9. Internal Multiplexers Functional Diagram ______________________________________________________________________________________ 19
Notebook CPU Step-Down Controller MAX1718T
+5V VCC
MAX1718T
3.0V TO 5.5V 26k 26k 26k 26k 26k
D4 100k 100k D3 D2 D1 D0 B-DATA LATCH
8k
8k
8k
8k
8k
GND
Figure 10. Internal Mux Impedance-Mode Data Test and Latch
resistors with the CPU's VID pins, each pin provides the low impedance needed for the mux to correctly interpret the Impedance mode setting. Unfortunately, the low resistances cause several mA quiescent currents for each of the CPU's grounded VID pins. This quiescent current can be avoided by taking advantage of the fact that D0-D4 need only appear low impedance briefly, not necessarily on a continuous DC basis. Highimpedance pullups can be used if they are bypassed with a large enough capacitance to make them appear low impedance for the 4s sampling interval. As noted in Figure 13, 4.7nF capacitors allow the inputs to appear low impedance even though they are pulled up with large-value resistors. Each sampling depletes some charge from the 4.7nF capacitors. A minimum interval of 2 R PULLUP 4.7nF is recommended between ZMODE samples. In some cases, it is desirable to determine the Impedance mode code during system boot so that sev-
3.0V TO 5.5V
D4 100k D3 ZMODE LOW VID = 01100 1.15V D2 100k D1 D0 ZMODE ZMODE HIGH VID = 01010 1.25V ZMODE = HIGH = 1.25V ZMODE = LOW = 1.15V
MAX1718T
Figure 11. Using the Internal Mux with Hardwired Logic-Mode and Impedance-Mode DAC Codes 20 ______________________________________________________________________________________
Notebook CPU Step-Down Controller MAX1718T
2.7V TO 5.5V
1k
100k
D4
MAX1718T
D3 D2 D1 D0 ZMODE 1k 100k 1k ZMODE = HIGH = 1.25V ZMODE = LOW = 1.15V
NOTE: USE PULLUP FOR LOGIC MODE 1, PULL-DOWN FOR LOGIC MODE 0. USE 100k FOR IMPEDANCE MODE 1, 1k FOR IMPEDANCE MODE 0.
Figure 12. Using the Internal Mux with Both VID Codes Resistor Programmed
eral processor types can be used without hardware modifications. Figure 14 shows one way to implement this function. The desired code is determined by the system BIOS and programmed into one register of the MAX1609 using the SMBusTM serial interface. The MAX1609's other register is left in its power-up state (all outputs high impedance). When SMBSUS is low, the outputs are high impedance and do not affect the logicmode VID code setting. When SMBSUS is high, the programmed register is selected, and the MAX1609 forces a low impedance on the appropriate VID input pins. The ZMODE signal is delayed relative to the SMBSUS pin because the VID pins that are pulled low by the MAX1609 take significant time to rise when they are released. One additional benefit of using the MAX1609 for this application is that the application uses only five of the MAX1609's high-voltage, open-drain outputs. The other three outputs can be used for other purposes.
Table 5. DAC Mux Operation
ZMODE GND VCC X SUS GND GND VCC OUTPUT VOLTAGE DETERMINED BY Logic level of D0-D4 Impedance of D0-D4 Logic levels of S0, S1
age level with the lowest possible peak currents for a given output capacitance. This makes the IC ideal for mobile CPUs. Mobile CPUs operate at multiple clock frequencies, which often require distinct VID settings. When transitioning from one clock frequency to the other, the CPU first goes into a low-power state, then the output voltage and clock frequency are changed. The change must be accomplished in a specified transition time or the system can halt. At the beginning of an output voltage transition, the MAX1718T blanks the VGATE output, preventing it from going low. VGATE remains blanked during the transition and is reenabled when the slew-rate controller has set the internal DAC to the final value and one addition-
Output Voltage Transition Timing
The MAX1718T is designed to perform output voltage transitions in a controlled manner, automatically minimizing input surge currents. This feature allows the circuit designer to achieve nearly ideal transitions, guaranteeing just-in-time arrival at the new output voltSMBus is a trademark of Intel Corp.
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Notebook CPU Step-Down Controller MAX1718T
3.15V TO 5.5V *OPTIONAL
4.7nF
1M
1k
1k
1k
1k
1k
D4 100k D3 CPU 100k D1 D0 ZMODE CPU VID = 01100 1.15V (ZMODE LOW) ZMODE HIGH VID = 01010 1.25V ZMODE = HIGH = 1.25V ZMODE = LOW = 1.15V D2
MAX1718T
*TO REDUCE QUIESCENT CURRENT, 1k PULLUP RESISTORS CAN BE REPLACED BY 1M RESISTORS WITH 4.7nF CAPACITORS IN PARALLEL.
Figure 13. Using the Internal Mux with CPU Driving the Logic-Mode VID Code
al slew-rate clock period has passed. The slew-rate clock frequency (set by resistor RTIME) must be set fast enough to ensure that the longest required transition is completed within the allowed interval. The output voltage transition is performed in 25mV steps, preceded by a 4s delay and followed by one additional clock period. The total time for a transition depends on R TIME , the voltage difference, and the accuracy of the MAX1718T's slew-rate clock, and is not dependent on the total output capacitance. The greater the output capacitance, the higher the surge current required for the transition. The MAX1718T automatically controls the current to the minimum level required to complete the transition in the calculated time, as long as the surge current is less than the current limit set by ILIM. The transition time is given by: 1 VOLD - VNEW 4s + fSLEW 25mV
where fSLEW = 150kHz 120k / RTIME, VOLD is the original DAC setting, and VNEW is the new DAC setting. See Time Frequency Accuracy in the Electrical Characteristics table for fSLEW accuracy. The practical range of RTIME is 47k to 470k, corresponding to 2.6s to 26s per 25mV step. Although the DAC takes discrete 25mV steps, the output filter makes the transitions relatively smooth. The average inductor current required to make an output voltage transition is: IL COUT 25mV fSLEW
Output Overvoltage Protection
The overvoltage protection (OVP) circuit is designed to protect the CPU against a shorted high-side MOSFET by drawing high current and blowing the battery fuse. The output voltage is continuously monitored for overvoltage. If the output is more than 2V, OVP is triggered and the circuit shuts down. The DL low-side gate-driver output is then latched high until SKP/SDN is toggled or VCC power is cycled below 1V. This action turns on the synchronous-rectifier MOSFET with 100% duty and, in turn, rapidly discharges the output filter capacitor and
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Notebook CPU Step-Down Controller MAX1718T
3.3V
R
R
R
R
R = 100k R VID4 R VID3
CPU
R VID2 R VID1 R VID0 ZMODE 3.3V MAX1718T
MAX1609 ADD0 ADDRESS ADD1 0 1 1 0 1 1 1 DATA SMBUS CLOCK 0 1 1 SMBSUS
3.3k
1nF
GMUXSEL
Figure 14. Using the ZMODE Multiplexer
forces the output to ground. If the condition that caused the overvoltage (such as a shorted high-side MOSFET) persists, the battery fuse blows. DL is also kept high continuously when VCC UVLO is active, as well as in shutdown mode (Table 6). Overvoltage protection can be defeated with a logic high on OVP or through the NO FAULT test mode (see the NO FAULT Test Mode section).
restart until VCC power is cycled or SKP/SDN is toggled. To allow startup, UVP is ignored during the undervoltage fault-blanking time (the first 256 cycles of the slew rate after startup). UVP can be defeated through the NO FAULT test mode (see the NO FAULT Test Mode section).
NO FAULT Test Mode
The over/undervoltage protection features can complicate the process of debugging prototype breadboards since there are (at most) a few milliseconds in which to determine what went wrong. Therefore, a test mode is provided to disable the OVP, UVP, and thermal shutdown features, and clear the fault latch if it has been
23
Output Undervoltage Shutdown
The output UVP function is similar to foldback current limiting, but employs a timer rather than a variable current limit. If the MAX1718T output voltage is under 70% of the nominal value, the PWM is latched off and does not
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Notebook CPU Step-Down Controller MAX1718T
Table 6. Operating Mode Truth Table
SKP/SDN GND DL High MODE Shutdown COMMENT Low-power shutdown state. DL is forced to VDD, enforcing OVP. ICC + IDD = 2A (typ). Test mode with faults disabled and fault latches cleared, including thermal shutdown. Otherwise, normal operation, with automatic PWM/PFM switchover for pulse skipping at light loads. Low-noise operation with no automatic switchover. Fixed-frequency PWM action is forced regardless of load. Inductor current reverses at light-load levels. Operation with automatic PWM/PFM switchover for pulse skipping at light loads. Fault latch has been set by OVP, UVP, or thermal shutdown. Device remains in FAULT mode until VCC power is cycled or SKP/SDN is forced low.
12V to 15V
Switching
No fault
Open
Switching
Run (PWM, low noise)
VCC
Switching
Run (PFM/PWM)
VCC or open
High
Fault
set. The PWM operates as if SKP/SDN were high (SKIP mode). The NO FAULT test mode is entered by forcing 12V to 15V on SKP/SDN.
Design Procedure
Firmly establish the input voltage range and maximum load current before choosing a switching frequency and inductor operating point (ripple-current ratio). The primary design trade-off lies in choosing a good switching frequency and inductor operating point, and the following four factors dictate the rest of the design: 1) Input Voltage Range. The maximum value (VIN(MAX)) must accommodate the worst-case high-AC adapter voltage. The minimum value (VIN(MIN)) must account for the lowest battery voltage after drops due to connectors, fuses, and battery selector switches. If there is a choice at all, lower input voltages result in better efficiency. 2) Maximum Load Current. There are two values to consider. The peak load current (ILOAD(MAX)) determines the instantaneous component stresses and filtering requirements, and thus drives output capacitor selection, inductor saturation rating, and the design of the current-limit circuit. The continuous load current (ILOAD) determines the thermal stresses and thus drives the selection of input capacitors, MOSFETs, and other critical heat-contributing components. Modern notebook CPUs generally exhibit ILOAD = ILOAD(MAX) 80%. 3) Switching Frequency. This choice determines the basic trade-off between size and efficiency. The optimal frequency is largely a function of maximum input voltage, due to MOSFET switching losses that are pro24
portional to frequency and VIN2. The optimum frequency is also a moving target, due to rapid improvements in MOSFET technology that are making higher frequencies more practical. 4) Inductor Operating Point. This choice provides tradeoffs between size and efficiency. Low inductor values cause large ripple currents, resulting in the smallest size, but poor efficiency and high output noise. The minimum practical inductor value is one that causes the circuit to operate at the edge of critical conduction (where the inductor current just touches zero with every cycle at maximum load). Inductor values lower than this grant no further size-reduction benefit. The MAX1718T's pulse-skipping algorithm initiates skip mode at the critical conduction point. So, the inductor operating point also determines the loadcurrent value at which PFM/PWM switchover occurs. The optimum point is usually found between 20% and 50% ripple current. 5) The inductor ripple current also impacts transientresponse performance, especially at low VIN - VOUT differentials. Low inductor values allow the inductor current to slew faster, replenishing charge removed from the output filter capacitors by a sudden load step. The amount of output sag is also a function of the maximum duty factor, which can be calculated from the on-time and minimum off-time:
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Notebook CPU Step-Down Controller
V (I LOAD1 - I LOAD2 )2 x L K OUT + t OFF(MIN) VIN V - VOUT 2 x COUT x VOUT K IN - t OFF(MIN) VIN
ILIMIT(LOW) = 95mV / 5.7m = 16.7A and the required valley current limit is: ILIMIT(LOW) > 19A - (0.30 / 2) 19A = 16.2A Since 16.7A is greater than the required 16.2A, the circuit can deliver the full-rated 19A. When delivering 19A of output current, the worst-case power dissipation of Q2 is 1.95W. With a thermal resistance of +60C/W and each MOSFET dissipating 0.98W, the temperature rise of the MOSFETs is +60C/W 0.98W = +58C, and the maximum ambient temperature is +125C - +58C = +67C. To operate at a higher ambient temperature, choose lower RDS(ON) MOSFETs or reduce the thermal resistance. Raising the current-limit threshold allows for operation with a higher MOSFET junction temperature. Connect ILIM to VCC for a default 100mV current-limit threshold. For an adjustable threshold, connect a resistordivider from REF to GND, with ILIM connected to the center tap. The external adjustment range of 0.5V to 3.0V corresponds to a current-limit threshold of 50mV to 300mV. When adjusting the current limit, use 1% tolerance resistors and a 10A divider current to prevent a significant increase of errors in the current-limit tolerance.
MAX1718T
VSAG =
where t OFF(MIN) is the minimum off-time (see the Electrical Characteristics tables) and K is from Table 2.
Inductor Selection
The switching frequency and operating point (% ripple or LIR) determine the inductor value as follows: L= VOUT VIN - VOUT
VIN x fSW x LIR x ILOAD(MAX)
(
)
Example: ILOAD(MAX) = 19A, VIN = 7V, VOUT = 1.25V, fSW = 300kHz, 30% ripple current or LIR = 0.30: VSOAR L x I PEAK 2 2 x C x VOUT
Find a low-loss inductor having the lowest possible DC resistance that fits in the allotted dimensions. Ferrite cores are often the best choice, although powdered iron is inexpensive and can work well at 200kHz. The core must be large enough not to saturate at the peak inductor current (IPEAK): IPEAK = ILOAD(MAX) + (LIR / 2) ILOAD(MAX)
Output Capacitor Selection
The output filter capacitor must have low enough effective series resistance (ESR) to meet output ripple and loadtransient requirements, yet have high enough ESR to satisfy stability requirements. Also, the capacitance value must be high enough to absorb the inductor energy going from a full-load to no-load condition without tripping the OVP circuit. In CPU VCORE converters and other applications where the output is subject to violent load transients, the output capacitor's size typically depends on how much ESR is needed to prevent the output from dipping too low under a load transient. Ignoring the sag due to finite capacitance: RESR VSTEP / ILOAD(MAX) The actual microfarad capacitance value required often relates to the physical size needed to achieve low ESR, as well as to the chemistry of the capacitor technology. Thus, the capacitor is usually selected by ESR and voltage rating rather than by capacitance value (this is true of tantalums, OS-CONs, and other electrolytics). When using low-capacity filter capacitors such as ceramic or polymer types, capacitor size is usually determined by the capacity needed to prevent VSAG
25
Setting the Current Limit
The minimum current-limit threshold must be great enough to support the maximum load current when the current limit is at the minimum tolerance value. The valley of the inductor current occurs at ILOAD(MAX) minus half of the ripple current; therefore: ILIMIT(LOW) > ILOAD(MAX) - (LIR / 2) ILOAD(MAX) where I LIMIT(LOW) equals the minimum current-limit threshold voltage divided by the RDS(ON) of Q2. For the MAX1718T Figure 1 circuit, the minimum current-limit threshold with VILIM = 105mV is about 95mV. Use the worst-case maximum value for RDS(ON) from the MOSFET Q2 data sheet, and add some margin for the rise in RDS(ON) with temperature. A good general rule is to allow 0.5% additional resistance for each C of temperature rise. Examining the Figure 1 example with a Q2 maximum RDS(ON) = 3.8m at TJ = +25C and 5.7m at TJ = +125C reveals the following:
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Notebook CPU Step-Down Controller MAX1718T
and VSOAR from causing problems during load transients. Generally, once enough capacitance is added to meet the overshoot requirement, undershoot at the rising load edge is no longer a problem (see the VSAG equation in the Design Procedure section). The amount of overshoot due to stored inductor energy can be calculated as: L x I PEAK 2 VSOAR 2 x C x VOUT where IPEAK is the peak inductor current. MAX1718T is operated as the second stage of a twostage power-conversion system, tantalum input capacitors are acceptable. In either configuration, choose an input capacitor that exhibits less than +10C temperature rise at the RMS input current for optimal circuit longevity.
Power MOSFET Selection
Most of the following MOSFET guidelines focus on the challenge of obtaining high load-current capability (>12A) when using high-voltage (>20V) AC adapters. Low-current applications usually require less attention. The high-side MOSFET must be able to dissipate the resistive losses plus the switching losses at both VIN(MIN) and VIN(MAX). Calculate both of these sums. Ideally, the losses at VIN(MIN) should be roughly equal to the losses at VIN(MAX), with lower losses in between. If the losses at VIN(MIN) are significantly higher than the losses at VIN(MAX), consider increasing the size of Q1. Conversely, if the losses at VIN(MAX) are significantly higher than the losses at VIN(MIN), consider reducing the size of Q1. If VIN does not vary over a wide range, the minimum power dissipation occurs where the resistive losses equal the switching losses. Choose a low-side MOSFET (Q2) that has the lowest possible RDS(ON), comes in a moderate-sized package (i.e., two or more 8-pin SOs, DPAKs, or D2PAKs), and is reasonably priced. Ensure that the MAX1718T DL gate driver can drive Q2; in other words, check that the dv/dt caused by Q1 turning on does not pull up the Q2 gate due to drain-to-gate capacitance, causing crossconduction problems. Switching losses are not an issue for the low-side MOSFET since it is a zero-voltage switched device when used in the buck topology.
Output Capacitor Stability Considerations
Stability is determined by the value of the ESR zero relative to the switching frequency. The voltage-positioned circuit in this data sheet has the ESR zero frequency lowered due to the external resistor in series with the output capacitor ESR, guaranteeing stability. For a voltage-positioned circuit, the minimum ESR requirement of the output capacitor is reduced by the voltage-positioning resistor value. The boundary condition of instability is given by the following equation: (RESR + RDROOP) COUT 1 / (2 fSW) where RDROOP is the effective value of the voltage-positioning resistor (Figure 1, R8). For good phase margin, increase the equivalent RC time constant by a factor of two. The standard application circuit (Figure 1) operating at 300kHz with COUT = 1320F, RESR = 2.5m, and RDROOP = 5m easily meets this requirement. In some applications, the COUT and RDROOP values are sufficient to guarantee stability even if RESR = 0. The easiest method for checking stability is to apply a very fast zero-to-max load transient and carefully observe the output voltage ripple envelope for overshoot and ringing. Do not allow more than one cycle of ringing after the initial step-response under/overshoot.
MOSFET Power Dissipation
The high-side MOSFET power dissipation due to resistance is: V PD (Q1 Re sistive) = OUT x I LOAD2 x RDS(ON) VIN Generally, a small high-side MOSFET is desired to reduce switching losses at high input voltages. However, the RDS(ON) required to stay within package power-dissipation limits often limits how small the MOSFET can be. Switching losses in the high-side MOSFET can become an insidious heat problem when maximum AC adapter voltages are applied, due to the squared term in the CV2fSW switching-loss equation. If the high-side MOSFET chosen for adequate RDS(ON) at low battery voltages
Input Capacitor Selection
The input capacitor must meet the ripple current requirement (IRMS) imposed by the switching currents defined by the following equation: I RMS = I LOAD VOUT (VIN - VOUT ) VIN
For most applications, nontantalum chemistries (ceramic or OS-CON) are preferred due to their resistance to inrush surge currents typical of systems with a switch or a connector in series with the battery. If the
26
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Notebook CPU Step-Down Controller
becomes extraordinarily hot when subjected to VIN(MAX), reconsider your choice of MOSFET. Calculating the power dissipation in Q1 due to switching losses is difficult since it must allow for difficult quantifying factors that influence the turn-on and turnoff times. These factors include the internal gate resistance, gate charge, threshold voltage, source inductance, and PC board layout characteristics. The following switching-loss calculation provides only a very rough estimate and is no substitute for breadboard evaluation and temperature measurements: PD(Q1 Switching) = CRSS x VIN(MAX)2 x fSW x I LOAD IGATE
Applications Information
Voltage Positioning
Powering new mobile processors requires new techniques to reduce cost, size, and power dissipation. Voltage positioning reduces the total number of output capacitors to meet a given transient response requirement. Setting the no-load output voltage slightly higher allows a larger step down when the output current suddenly increases, and regulating at the lower output voltage under load allows a larger step up when the output current suddenly decreases. Allowing a larger step size means that the output capacitance can be reduced and the capacitor's ESR can be increased. Adding a series output resistor positions the full-load output voltage below the actual DAC programmed voltage. Connect FB directly to the inductor side of the voltagepositioning resistor (R8, 4m). The other side of the voltage-positioning resistor should be connected directly to the output filter capacitor with a short, wide PC board trace. With a 20A full-load current, R8 causes an 80mV drop. This 80mV is a -6.4% droop. An additional benefit of voltage positioning is reduced power consumption at high load currents. Because the output voltage is lower under load, the CPU draws less current. The result is lower power dissipation in the CPU, although some extra power is dissipated in R8. For a nominal 1.25V, 20A output, reducing the output voltage 6.4% gives an output voltage of 1.17V and an output current of 18.7A. Given these values, CPU power consumption is reduced from 25W to 21.9W. The additional power consumption of R8 is: 4m 18.7A2 = 1.4W And the overall power savings is as follows: 25 - (21.9 + 1.4) = 1.7W In effect, 3W of CPU dissipation is saved, and the power supply dissipates some of the power savings, but both the net savings and the transfer of dissipation away from the hot CPU are beneficial.
MAX1718T
where CRSS is the reverse transfer capacitance of Q1 and IGATE is the peak gate-drive source/sink current (2A typ). For the low-side MOSFET (Q2), the worst-case power dissipation always occurs at maximum battery voltage: VOUT 2 PD (Q2) = 1 - I LOAD x RDS(ON) VIN(MAX) For both Q1 and Q2, note the MOSFET's maximum junction temperature and the thermal resistance that is realistically achieved with the device packaging and your thermal environment to avoid overheating. The absolute worst case for MOSFET power dissipation occurs under heavy overloads that are greater than ILOAD(MAX) but are not quite high enough to exceed the current limit and cause the fault latch to trip. To protect against this possibility, you can "overdesign" the circuit to tolerate: ILOAD = ILIMIT(HIGH) + (LIR / 2) ILOAD(MAX) where I LIMIT(HIGH) is the maximum valley current allowed by the current-limit circuit, including threshold tolerance and on-resistance variation. This means that the MOSFETs must be very well heatsinked. If short-circuit protection without overload protection is enough, a normal ILOAD value can be used for calculating component stresses. Choose a Schottky diode (D1) having a forward voltage low enough to prevent the Q2 MOSFET body diode from turning on during the dead time. As a general rule, a diode having a DC current rating equal to 1/3 of the load current is sufficient. This diode is optional and can be removed if efficiency is not critical.
Reduced-Power-Dissipation Voltage Positioning
A key benefit of voltage positioning is reduced power dissipation, especially at heavy loads. In the standard application circuit (Figure 1), voltage positioning is accomplished using a droop resistor (R8), which can dissipate over 1W. Although the power savings in the processor is much greater than the dissipation in the resistor, 1W of dissipation is still far from ideal.
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Notebook CPU Step-Down Controller
The resistor is a necessary component because accurate voltage positioning depends on an accurate current-sense element. But it is not necessary to drop the entire positioning voltage across this resistor. The circuit of Figure 15 uses an external op amp to add gain to R8's voltage signal, allowing the resistor value and power dissipation to be divided by the gain factor. The recommended range for the gain is up to about 4, with preferred practical values around 1.5-3. There are several difficulties with high gains. If high gain is used, the sense-resistor value is very small (<1m). The sense signal is also small, potentially causing noise and stability problems. Also, output voltage and positioning accuracy are essential. A smaller sense signal reduces accuracy, as does any op-amp input voltage offset, which is increased by the gain factor. The op-amp output directly drives FB. To ensure stability, the output voltage ripple and the ripple signal across R8 must be delivered with good fidelity. To preserve higher harmonics in the ripple signal, the circuit bandwidth should be approximately 10 times the switching frequency. Besides lowering power dissipation, the gain stage provides another benefit; it eases the task of providing the required positioning slope, using available discrete values for R8. A lower value resistor can be used and the gain adjusted to deliver the desired slope. Sometimes the desired slope is not well known before the final PC board is evaluated. A good practice is to adjust the final gain to deliver the correct voltage slope at the processor pins, adjusting for the actual copper losses in the supply and ground paths. This does not remove the requirement to minimize copper losses because they vary with temperature and PC board production lot. It does, however, provide an easy, practical way to account for their typical expected voltage drops. Replacing the droop resistor with a lower value resistor and a gain stage does not affect the MAX1718T stability criteria. The IC cannot distinguish one from the other, as long as the required signal integrity is maintained. R8's effective value can be used to guarantee stability with extremely low-ESR (ceramic) output capacitors (see the Output Capacitor Stability Considerations section). The effective value is the resistor value that would result in the same signal delivered to the MAX1718T's FB pin, or R8 times the op-amp circuit's gain. Although the op amp should be placed near R8 to minimize input noise pickup, power it from the MAX1718T's quiet VCC supply and analog ground to prevent other noise problems.
MAX1718T
DH
R8
VOUT 1.25V, 19A
DL MAX1718T
VCC 510
FB A=2 R5 1k
MAX4322
R6 1k
Figure 15. Lowering Voltage-Positioning Power Dissipation
High-Accuracy Current Limit
The MAX1718T's integrated current limit uses the synchronous rectifier's RDS(ON) for its current-sense element. This dependence on a poorly specified resistance with high temperature variation means that the integrated current limit is useful mainly in high-overload and short-circuit conditions. A moderate overload may be tolerated indefinitely. This arrangement is tolerable because there are other ways to detect overload conditions and take appropriate action. For example, if the CPU draws excessive current (but not enough to activate the current limit) the CPU heats up, and eventually the system takes notice and shuts down. While this approach is usually acceptable, it is far from optimal. An inaccurate current limit causes component specification difficulties. What values should be used for inductor saturation ratings, MOSFET peak current requirements, and power dissipation requirements? An accurate current limit makes these issues more manageable. The circuit of Figure 16 uses an external op amp, together with the voltage-positioning resistor (R8) to implement an accurate inductor current limit. A voltage-divider from the positive side of R8 creates a threshold several mVs below the output. When the voltage drop across R8 exceeds the threshold, current limiting occurs. The op amp causes current limiting by lowering the voltage on the ILIM pin. This lowers the current-limit threshold of the IC's internal current-limit circuit, which uses the MOSFET RDS(ON) as usual. The op-amp output swing has the ability to adjust the IC's internal valley current limit from a value much higher than ever needed (given the MOSFET's RRD(ON)) to a
28
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Notebook CPU Step-Down Controller
value much lower than required to support a normal load. The bandwidth of the ILIM pin is not high, so the speed of the op amp is not critical. Any op amp or comparator could be acceptable, as long as its input offset does not degrade current-limit accuracy excessively, has input common-mode range to ground, and has Rail-toRail(R) output swing. Because the bandwidth is low, the circuit responds to the average inductor current rather than the peak or valley current, eliminating the current limit's dependence on inductor ripple current. Similar to a foldback current limit, this circuit must be carefully designed to guarantee startup. The op amp must be incapable of setting the current limit to zero or else the power supply may be unable to start. The three-way divider from REF to ground to the op-amp output allows the op amp to vary the MAX1718T's internal current-limit threshold from 21mV (severely limiting current) to 182mV (more than guaranteeing the maximum required output current). These divider resistors should be chosen with the required current and the synchronous rectifier's RRD(ON) in mind to ensure that the op-amp adjustment range is high enough to guarantee the required output current. The voltage at the ILIM pin is given by: VILIM =
MAX1718T
DH
R8 4m
VOUT 1.25V, 19A
DL MAX1718T FB VCC REF R14 100k R12 30k ILIM 1nF R13 20k MAX4322 R11 20k R10 1.5k
Figure 16. Improving Current-Limit Accuracy
output capacitor may be large enough to require the current limit to be increased beyond what is necessary to support the load. This circuit is completely compatible with the circuit of Figure 15. If the two circuits are used together, the MAX4326 dual op amp in a MAX package can replace the two single devices, saving space and cost. If both are used, the reduced R8 value makes the opamp input offset more significant. Additional margin might be needed, depending on the magnitude of R8's reduction. Although the op amp should be placed near R8 to minimize input noise pickup, power it from the MAX1718T's quiet VCC supply and analog ground to prevent other noise problems.
[(R12 x R13 x VREF ) + (R13 x R14 x VCOMP )] [(R12 x R14) + (R12 x R13) + (R13 x R14)]
where VCOMP is the voltage at the output of the comparator. The minimum VILIM is calculated when VCOMP is at the VOL of the comparator. The maximum VILIM is calculated when VCOMP = VOH at the minimum VCC. The valley current-limit threshold is set at 10% of the voltage VILIM. C13 should be picked to give approximately 10s time constant at the ILIM input. The actual threshold at which the op amp begins to limit current is determined by R8 and the R10/R11 divider values and is very easy to set. Ideally, IOUT(MAX) R8 = VFB R10 / (R10 + R11). In practice, some margin must be added for resistor accuracy, opamp input offset, and general safety. With the op amp shown and 1% resistors, 10% margin is adequate. An additional benefit of this circuit is that the currentlimit value is proportional to the output voltage setting (VFB). When the output voltage setting is lowered, the current limit automatically adjusts to a more appropriate level, providing additional protection without compromising performance since the reduction of the required load current is greater than that of the output voltage setting. In some cases, the current required to slew the
Rail-to-Rail is a registered trademark of Nippon Motorola, Ltd.
Using Skip Mode During Suspend (SKP/SDN = VCC)
Typically, for the MAX1718T's intended application, the minimum output currents are too high to benefit from pulse-skipping operation in all active CPU modes. Furthermore, skip mode can be a hindrance to properly executing downward output voltage transitions (see the Forced-PWM Mode section). However, processor suspend currents can be low enough that skip mode operation provides a real benefit. In the circuit of Figure 17, SKP/SDN remains biased at 2V in every state except suspend and shutdown. In addition, upon entering suspend (SUS going high), the pin remains at 2V for about 200s before it eventually
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29
Notebook CPU Step-Down Controller MAX1718T
goes high. This causes the MAX1718T to remain in PWM mode long enough to correctly complete the negative output voltage transition to the suspend state voltage. When SKP/SDN goes high, the MAX1718T enters its low-quiescent-current skip mode. ed, calculate VSAG to be sure of adequate transient response. Dropout Design Example: VOUT = 1.6V fsw = 550kHz K = 1.8s, worst-case K = 1.58s TOFF(MIN) = 500ns VDROP1 = VDROP2 = 100mV h = 1.5 VIN(MIN) = (1.6V + 0.1V) / (1-0.5s 1.5/1.58s) + 0.1V - 0.1V = 3.2V Calculating again with h = 1 gives the absolute limit of dropout: VIN(MIN) = (1.6V + 0.1V) / (1-1.0 0.5s/1.58s) - 0.1V + 0.1V = 2.5V Therefore, VIN must be greater than 2.5V, even with very large output capacitance, and a practical input voltage with reasonable output capacitance would be 3.2V.
Dropout Performance
The output voltage adjust range for continuous-conduction operation is restricted by the nonadjustable 500ns (max) minimum off-time one-shot (375ns max at 1000kHz). For best dropout performance, use the slower (200kHz) on-time settings. When working with low input voltages, the duty-factor limit must be calculated using worst-case values for on- and off-times. Manufacturing tolerances and internal propagation delays introduce an error to the TON K-factor. This error is greater at higher frequencies (Table 2). Also, keep in mind that transient response performance of buck regulators operated close to dropout is poor, and bulk output capacitance must often be added (see the VSAG equation in the Design Procedure section). The absolute point of dropout is when the inductor current ramps down during the minimum off-time (IDOWN) as much as it ramps up during the on-time (IUP). The ratio h = IUP/IDOWN is an indicator of ability to slew the inductor current higher in response to increased load, and must always be greater than 1. As h approaches 1, the absolute minimum dropout point, the inductor current is less able to increase during each switching cycle and V SAG greatly increases unless additional output capacitance is used. A reasonable minimum value for h is 1.5, but this may be adjusted up or down to allow tradeoffs between V SAG , output capacitance, and minimum operating voltage. For a given value of h, the minimum operating voltage can be calculated as: VIN(MIN) =
Adjusting VOUT with a Resistor-Divider
The output voltage can be adjusted with a resistordivider rather than the DAC if desired (Figure 18). The drawback is that the on-time does not automatically receive correct compensation for changing output voltage levels. This can result in variable switching frequency as the resistor ratio is changed, and/or excessive switching frequency. The equation for adjusting the output voltage is: R VOUT = VFB 1 + 1 R2 where V FB is the currently selected DAC value. In resistor-adjusted circuits, the DAC code should be set as close as possible to the actual output voltage in order to minimize the shift in switching frequency.
TOFF(MIN ) x h 1- K
(VOUT + VDROP1)
+ VDROP2 - VDROP1
One-Stage (Battery Input) vs. Two-Stage (5V Input) Applications
The MAX1718T can be used with a direct battery connection (one stage) or can obtain power from a regulated 5V supply (two stage). Each approach has advantages, and careful consideration should go into the selection of the final design. The one-stage approach offers smaller total inductor size and fewer capacitors overall due to the reduced demands on the 5V supply. The transient response of the single stage is better due to the ability to ramp up the inductor current faster. The total efficiency of a single stage is better than the two-stage approach.
where VDROP1 and VDROP2 are the parasitic voltage drops in the discharge and charge paths, respectively (see the On-Time One-Shot (TON) section), TOFF(MIN) is from the Electrical Characteristics tables, and K is taken from Table 2. The absolute minimum input voltage is calculated with h = 1. If the calculated VIN(MIN) is greater than the required minimum input voltage, then operating frequency must be reduced or output capacitance added to obtain an acceptable VSAG. If operation near dropout is anticipat30
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Notebook CPU Step-Down Controller MAX1718T
VCC
VBATT DH
120k VCC 80k TO SUS 30k 0.01F TO SKP/SDN
VOUT DL
MAX1718T
FB
SHUTDOWN 3.3V
R1
R2
0V
SUS
~200s
~200s
5V
SKP/SDN
2.0V
Figure 17. Using Skip Mode During Suspend (SKP/ SDN = VCC)
VOUT = VFB 1 + R1 R2
(
)
The two-stage approach allows flexible placement due to smaller circuit size and reduced local power dissipation. The power supply can be placed closer to the CPU for better regulation and lower I2R losses from PC board traces. Although the two-stage design has worse transient response than the single stage, this can be offset by the use of a voltage-positioned converter.
Figure 18. Adjusting VOUT with a Resistor-Divider
Ceramic Output Capacitor Applications
Ceramic capacitors have advantages and disadvantages. They have ultra-low ESR and are noncombustible, relatively small, and nonpolarized. They are also expensive and brittle, and their ultra-low ESR characteristic can result in excessively high ESR zero frequencies (affecting stability in nonvoltage-positioned circuits). In addition, their relatively low capacitance value can cause output overshoot when going abruptly from full-load to no-load conditions, unless the inductor value can be made small (high-switching frequency), or there are some bulk tantalum or electrolytic capacitors in parallel to absorb the stored energy in the inductor. In some cases, there may be no room for electrolytics, necessitating a ceramic-only DC-to-DC design. The MAX1718T can take full advantage of the small size and low ESR of ceramic output capacitors in a voltage-positioned circuit. Adding the positioning resistor increases the ripple at FB, lowering the effective ESR zero frequency of the ceramic output capacitor. Output overshoot (VSOAR) determines the minimum output capacitance requirement (see the Output Capacitor Selection section). Often the switching frequency is increased to 550kHz or 1000kHz, and the inductor value is reduced to minimize the energy transferred from
inductor to capacitor during load-step recovery. The efficiency penalty for operating at 550kHz is about 2% to 3% and about 5% at 1000kHz when compared to the 300kHz voltage-positioned circuit, primarily due to the high-side MOSFET switching losses.
PC Board Layout Guidelines
Careful PC board layout is critical to achieve low switching losses and clean, stable operation. The switching power stage requires particular attention (Figure 19). If possible, mount all of the power components on the top side of the board with their ground terminals flush against one another. Follow these guidelines for good PC board layout: 1) Keep the high-current paths short, especially at the ground terminals. This is essential for stable, jitterfree operation. 2) All analog grounding is done to a separate solid copper plane, which connects to the MAX1718T at the GND pin. This includes the V CC , REF, and CC capacitors, the TIME resistor, as well as any other resistor-dividers. 3) Keep the power traces and load connections short. This is essential for high efficiency. The use of thick copper PC boards (2oz vs. 1oz) can enhance full-load efficiency by 1% or more. Correctly routing PC board traces is a difficult task that must be approached in terms of fractions of centimeters, where a single milliohm of excess trace resistance causes a measurable efficiency penalty.
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31
Notebook CPU Step-Down Controller
4) LX and GND connections to Q2 for current limiting must be made using Kelvin-sense connections to guarantee the current-limit accuracy. With 8-pin SO MOSFETs, this is best done by routing power to the MOSFETs from outside using the top copper layer, while connecting GND and LX inside (underneath) the 8-pin SO package. 5) When trade-offs in trace lengths must be made, it is preferable to allow the inductor charging path to be made longer than the discharge path. For example, it is better to allow some extra distance between the input capacitors and the high-side MOSFET than to allow distance between the inductor and the low-side MOSFET or between the inductor and the output filter capacitor. 6) Ensure the FB connection to the output is short and direct. In voltage-positioned circuits, the FB connection is at the junction of the inductor and the positioning resistor. 7) Route high-speed switching nodes away from sensitive analog areas (CC, REF, ILIM). Make all pin-strap control input connections (SKP/SDN, ILIM, etc.) to analog ground or VCC rather than power ground or VDD.
MAX1718T
This point must also be very close to the output capacitor ground terminal. 5) Connect the output power planes (VCORE and system ground planes) directly to the output filter capacitor positive and negative terminals with multiple vias. Place the entire DC-to-DC converter circuit as close to the CPU as is practical.
Pin Configuration
TOP VIEW
V+ 1 SKP/SDN 2 TIME 3 FB 4 NEG 5 CC 6 S0 7 S1 8 VCC 9 TON 10 REF 11 ILIM 12 POS 13 VGATE 14 28 DH 27 LX 26 BST 25 D0 24 D1
MAX1718T
23 D2 22 D3 21 D4 20 OVP 19 ZMODE 18 SUS 17 VDD 16 DL 15 GND
Layout Procedure
1) Place the power components first, with ground terminals adjacent (Q2 source, CIN-, COUT-, D1 anode). If possible, make all these connections on the top layer with wide, copper-filled areas. 2) Mount the controller IC adjacent to MOSFET Q2, preferably on the back side opposite Q2 in order to keep LX-GND current-sense lines and the DL drive line short and wide. The DL gate trace must be short and wide, measuring 10 to 20 squares (50 mils to 100 mils wide if the MOSFET is 1in from the controller IC). 3) Group the gate-drive components (BST diode and capacitor, VDD bypass capacitor) together near the controller IC. 4) Make the DC-to-DC controller ground connections as shown in Figure 19. This diagram can be viewed as having three separate ground planes: output ground, where all the high-power components go; the GND plane, where the GND pin and VDD bypass capacitors go; and an analog ground plane where sensitive analog components go. The analog ground plane and GND plane must meet only at a single point directly beneath the IC. These two planes are then connected to the high-power output ground with a short connection from GND to the source of the lowside MOSFET Q2 (the middle of the star ground).
QSOP
Chip Information
TRANSISTOR COUNT: 7190
32
______________________________________________________________________________________
Notebook CPU Step-Down Controller MAX1718T
VBATT GND IN
ALL ANALOG GROUNDS CONNECT TO LOCAL PLANE ONLY VIA TO GND NEAR Q2 SOURCE
MAX1718T
VCC CC Q1 REF VDD D1 Q2 COUT
CIN GND OUT
VOUT GND VIA TO SOURCE OF Q2 CONNECT LOCAL ANALOG GROUND PLANE DIRECTLY TO GND FROM THE SIDE OPPOSITE THE VDD CAPACITOR GND TO AVOID VDD GROUND CURRENTS FROM FLOWING IN THE ANALOG GROUND PLANE. VIA TO LX R6 L1 VIA TO FB AND FBS
NOTE: "STAR" GROUND IS USED. D1 IS DIRECTLY ACROSS Q2.
INDUCTOR DISCHARGE PATH HAS LOW-DC RESISTANCE
Figure 19. Power-Stage PC Board Layout Example
______________________________________________________________________________________
33
Notebook CPU Step-Down Controller MAX1718T
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)
QSOP.EPS
Note: The MAX1718T does not have a heat slug.
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
34 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2002 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.


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